Thursday, January 12, 2017

A low power PSK31 transmitter using a Class-E power amplifier and envelope modulation

Back in 1999, not too long after the first appearance of PSK31, I decided that I wanted to construct a beacon transmitter that would operate using this mode, but at the time the only practical means of generating PSK31 was with a computer, a sound card and an SSB transmitter.  Not wanting to tie up that much gear for this purpose I set about to use the PIC16C84 microcontroller which was popular among the homebrew builders at the time.

By this time the AM broadcast band had (relatively) recently been expanded up to 1705 kHz but very few stations occupied the new 1605-1705 kHz segment.  In perusing the FCC rules I noted that part 15 section 219 had been modified to allow low-power experimental operation in this new segment of the AM broadcast band and I decided that with the lack of activity in this frequency range that it was a good time to put up a "MedFER" (Medium Frequency Experimental Radio) beacon.
Figure 1:
The "Balanced Modulator" (Baseband) version of the PSK31
transmitter/exciter.  Built to test a concept, it has a few flaws,
but it did work.
Click on the image for a larger version.

The balanced modulator method

Upon investigating various methods of producing a PSK31 signal I experimented with the generation of a bipolar baseband signal that could be applied directly to a balanced mixer.  While this method worked well it had the problem than it required that all following stages be linear.

A diagram of the prototype of that transmitter may be seen in Figure 1.  For this transmitter a crystal-controlled oscillator is constructed using two transistors (Q1, Q2) and the output is buffered by U3, a 74HC00 quad NAND gate.  The frequency used for this circuit was unimportant as it was a "proof of concept" and I (think that I) used a 4.9152 MHz crystal which, although not in any amateur band, still allowed an "across the room" reception with a short length of wire as an antenna.  Following the first U3 NAND buffer the remaining sections are used to provide a two phase signal with the output split 180 degrees which fed a very simple balanced modulator consisting of just two diodes, a few capacitors and some resistors.

To provide modulation a PIC16C84 is used to provide a 32-step staircase modulation using PWM techniques.  This PWM output, done using "bit-bang" software uses a frequency of 1 kHz which is exactly 32 times that of PSK31's 31.25 Hz baseband frequency, and is then filtered with a two stage R/C low-pass filter network consisting first of a 4.7k resistor and 0.1uF capacitor followed by a second stage with a much higher impedance consisting of a 150k resistor and 0.033uF capacitor providing around 3dB of roll-off at the 31.25Hz baseband frequency and about 40dB of attenuation at the 1 kHz PWM rate and an acceptable amount of Inter-Symbol Interference ("ISI").  The result of this filtering is that the vast majority of the 1kHz energy is removed, leaving a pretty clean 31.25 Hz baseband signal.

Figure 2:
Phase diagram of balanced modulator
circuit in Figure 1.  The propagation
delay of the gates result in a rather
imprecise 180 degree phase shift
causing the upside-down "Vee"
in the phase diagram.
The filtered PWM output is then buffered and split into two signals, one of them inverted, using several op-amp sections and these two signals are applied differentially via simple R/C networks across the two diodes:  If the baseband signal from the PWM output were to go "positive" (e.g. above the mid-supply voltage)  the other side would go "negative" and turn on one diode, but it if were to swing the other way the other diode - fed with the RF signal that was 180 degrees out of phase with the first - would be turned on.  The end result is a fairly nice, linear BPSK envelope and baseband waveform when viewed on a receiver with an oscilloscope.

While it worked to prove a concept, this signal has a few shortcomings.  First, the RF signal from the oscillator and buffer is not likely to have a precise 50% duty cycle which means that a bit more RF energy would be available in one phase than the other, resulting in a somewhat "lopsided" BPSK amplitude envelope that only minimally affects demodulation and overall signal quality.  The other problem has to do with a NAND gate being used to provide the 180 degree phase shift (e.g. signal inversion) in that the addition of the inverting gate adds a few 10s of nanoseconds of propagation delay.  While this doesn't sound like much, it does amount to a significant number of degrees of phase even at low HF frequencies and the end result is that the "Phase Diagram" (see Figure 2) is slightly distorted and produces the inverted "vee" pattern.

While I could have gotten this method to work (e.g. used a bandpass/lowpass filter to get a nice, clean sine wave and a transformer to get the 180 degree phase shift) it does hove a down side:  All subsequent stages would need to be linear.  While not a great technical problem, it did mean that for the MedFER transmitter, which has a 100 milliwatt input power limit according to FCC rules, a linear final amplifier would have at best around 70% efficiency which would mean that I'd lose a bit more than 1dB of signal over an amplifier that was 100% efficient.  While this may not sound like much I figured that I could do better with a more efficient amplifier scheme.

The Amplitude Modulator Method

Having proven the ability to produce a reasonable quality PSK31 waveform with a lowly PIC I decided to try a different approach:  Apply high-level modulation to the output amplifier stage.  What's more, this amplifier stage need not be linear at all:  It could be a conventional Class C stage which would boost the efficiency to something around 80%, but I decided on going a step farther and use a Class-E amplifier.

Figure 3:
Diagram of the "AM" version of the transmitter using separate amplitude
and phase modulation paths, allowing a non-linear but highly efficient
Class-E output amplifier to be used.  The capacitor, diode and resistor
on the gate of Q1, the output transistor, are used to prevent the FET
from being stuck "on" and shorting out the power supply should
the RF drive disappear for any reason and the output of the NAND
gate driving it be left in a "high" state.
Click on the image for a larger version.
I first became aware of the Class-E amplifier more than a decade earlier when my friend Mark, WB7CAK, designed one for his LowFER (Low Frequency Experimental Radio) beacon that operated in the 160-190 kHz "experimenter's" band authorized by section 217 of FCC part 15.  As with MedFER operation, the input power was also limited - 1 watt in this case.  After a bit of number crunching and fiddling on the workbench Mark came up with a simple circuit and a few basic equations that described how such an amplifier could be built and published an article in the Western Update - a small publication tailored mostly for LowFER.  Because this publication may be difficult to find I have reproduced it with permission from the author and it may be found here:  (Link).

While the maths behind the derivation of the operation of a Class-E amplifier can be somewhat involved, the concept is quite simple:  When the drive signal to the transistor - typically a power MOSFET at LowFER frequencies - goes low, the transistor shuts off and it does this quickly (e.g. driven "hard") so that transistor spends as little time as possible "partially" conducting in between "on" and "off".  When the transistor turns off the voltage on the drain rises and is pulled up by the choke in the circuit, but then falls again due the "ringing" of a resonant circuit on the output tank.  Because this tank circuit is tuned appropriately, precisely at the time that the drain voltage hits zero the output transistor is switched back on.  The result of these two events is that the FET is either completely on or off which means that little or no power is dissipated in it and when the FET is (quickly!) turned back on, it does so just when the voltage swings to zero, practically eliminating any losses that would occur at that instant due to the intrinsic resistance of the FET absorbing the current and from other losses of components from tank circuit being "shorted out" when voltage is present.

Figure 4:
The constructed MedFER beacon transmitter, built on the bottom
of a weather resistant outdoor enclosure to be mounted at the base
of the antenna.
The result of all of this is an RF amplifier that (exclusive of the drive signal) is demonstrably capable of 95%-98% efficiency!  In the MedFER and LowFER world this means that with our power level being limited on the input, we will have, for all practical purposes, all of our input  power at our disposal rather than, say, 70-80% of it as would be the case with almost any other amplifier type.

The obvious problem with a Class-E amplifier is that the drive signal must be a fast rising/falling square-shaped wave that slams the transistor on and off which means that amplitude modulation of that drive signal is not easily managed if efficiency is to be maintained.  What one can do is to modulate the power supply feeding the amplifier instead.

Remembering that a PSK31 signal consists of two parts - the amplitude modulation and the phase shift - we can split these two signals in the modulator.  The first part, amplitude modulation,  may be done by modulating the supply voltage of the output amplifier stage.  The second part, phase modulation, may also be done early in the path of the drive signal simply by flipping the phase of the RF signal under computer control.  In order to keep the signal "clean" all we really need to do is to time the flipping of the phase with the amplitude being brought to zero so that we don't transmit the broadband "click" that would otherwise occur when we did this abrupt phase shift.  The schematic of the transmitter is depicted in Figure 3.

Figure 5:
The phase diagram of the signal
produced by the "Amplitude
Modulator" MedFER PSK31
beacon transmitter.  The phase
shift is precise and the intermodulation
products are well within the tolernaces
dictated by good operating practice.
In this circuit the frequency-determining crystal oscillator operates at four times the transmitter frequency, or around 6.8 MHz in the case of the MedFER transmitter.  During construction it was observed that at around 1.7 MHz it was was easier to achieve Class-E operation at this power level with a drive waveform that had a 25% duty cycle so a 74HC4017 counter was used, wired as a divide-by-four but giving two 25% duty cycle outputs, 180 degrees apart.  To select which of these signals were to be used a simple MUX and driver was constructed using four NAND gates, this time being designed so that the same amount of propagation delay would occur during either phase to eliminate the upside-down "Vee" seen in Figure 2.

The PWM signal was generated using simple R/C filtering in the same way as it was for the balanced modulator circuit, but this time op amps were used to set the offset and gain (or "span") so that the baseband waveform could be precisely adjusted in both amplitude and so that when the baseband signal went to zero, the output power from the Class-E circuit would as well, compensating for the voltage offset of the series modulating transistor, emitter-follower Q4.  The output transistor, Q3, is a low-power MOSFET wired into a simple L/C "tank" circuit that is tuned to result on the coincidence of the zero crossing of the drain voltage and the transistor being turned back on by the 25% duty cycle drive signal.  Multiple taps are provided on the tank coil making it easy to set both the output power and match it appropriately to the load presented by the resistance seen at the loading coil.
Figure 6:
Loading coil used to match the transmitter output to the
feedpoint impedance.  This coil is wound using 3/8"
copper tubing and uses a variometer inside the coil
to provide a low-loss means of adjusting the inductance.

For modulation the PIC produces a semi-sine waveform that looks very similar to one "cycle" on the double-frequency output of a full-wave diode rectifier and when this waveform amplitude is taken to "zero" another output of the PIC causes a phase switch to occur.  It is in this way that the BPSK modulation is broken into two parts - the phase change and the modulation envelope - and we are able to use a highly efficient, non-linear amplifier for the output.

After constructing this I later learned that a similar scheme was applied to  OSCAR amateur satellites (starting with OSCAR 7) that included linear transponders.  In order conserve precious power, the linear transponders were constructed using the "HELAPS" (High Efficiency Linear Amplifier using Parametric Synthesis) system where the amplitude and phase components of multiple signals in the satellite's passband were converted into their phase an amplitude components allowing both energy-saving class-C RF amplifiers and DC-DC switching converters to be used, the end result being a faithful, amplified reproduction of the input signal with a lower power budget that would have otherwise been required. This system was proposed by Dr. Karl Meinzer, DJ4ZC, and you can read about it on the AMSAT.DL web site here - link.

Where is it now?

This beacon was mounted in its enclosure on the roof of my house in 1999 and a rather large loading coil (see Figure 6) was to match its output impedance to the top-hatted 3 meter vertical antenna  - and it is there to this day.  While not regularly used, it still works, provided that the tuning of the loading coil is checked before operation.  Since the beacon was constructed more broadcast stations have taken to the air in the "new" AM segment, but its operating frequency - nominally 1704.965 kHz - is just below the top edge of the band, as far away from QRM as is possible.

In the past the BPSK31 signal from this beacon has been copied during the daylight hours at a distance of 75 air miles (approx. 120km) and it had been copied in various places in the western U.S. at night.  This beacon has since been modified to so that it may be externally on-off keyed so that "QRSS" (low-speed Morse with multi-second "dit" lengths) could be sent in addition to PSK31 allowing even greater distances to be spanned under more diverse conditions.
 
I haven't done much with the code for this transmitter other than add a few features when it was ported to the (then) newer PIC16F84.  Needless to say, there are more modern devices available that contain hardware that would have simplified the design such as that to generate a much higher frequency and higher resolution PWM signal and perhaps one day I'll investigate their use.

For more information on this and related projects - including schematics, various applications, more pictures and some source code, visit the "CT Medfer Beacon" web page - link and related pages linked from there.

[End]

Saturday, December 31, 2016

A simple push-pull audio amplifier using russian rod tubes and power transformers

As one sometimes does, I was perusing EvilBay a while back and saw some ex-USSR sub-miniature pentode tubes for sale.  In looking up the part number - 1Ж18Б, which is usually translated to "1J18B" (or perhaps "1Zh18B") I was intrigued as they were not "normal" tubes.

Many years ago I'd read about the type of tube that is now often referred to as a "Gammatron" - a "gridless" amplifier tube of the 1920s, so-designed to get around patents that included what would seem to be fundamental aspects of any tube such as the control grid.  Instead of a grid, the "third" control element was located near the "cathode" and "anode" - or even a pair of anodes.  As you might expect the effective gain of this type of tube was rather low and despite its working, it really didn't catch on.  It was the similarity between the description of the "Gammatron" and these "rod" tubes that interested me.
Figure 1:
A close-up of a 1J18B tube.  Note that the internals are a collection of rods
rather than "conventional" grids and plates.
Click on the image for a larger version.

Some information on the "Gammatron" tube - not to be confused with the later-used "Gammatron" product name - may be found at:
  • The Radio Museum - link.
  • The N6JV virtual tube museum - link.

In reading about these peculiar "rod" tubes I became intrigued, particularly after reading some threads about these tubes on the "radicalvalves" web site (link here) and the "radiomuseum" site (that link here).  Since they were pretty cheap I ordered some from a seller located in the former Soviet Union.

This past holiday week I managed to get a bit of spare time and decided to kludge together a simple circuit with some of these tubes which are pentodes with the suppressor grid internally connected to one side of the filament.  The first circuit was a simple, single-ended amplifier with one of these tubes wired as a triode.  Encouraged that it (kind of) worked I decided to put together a simple push-pull amplifier for more power.
Figure 2:
Diagram of the push-pull amplifier using 1J18B tubes wired as triodes.  On T1, a single 5 volt winding is
the audio input and the series 120 volt primaries, wired as if for a 240 volt connection, is used as a center-tapped winding
for the 180 degree split to feed the two tubes.  The speaker is connected to the "115" and "125" volt taps of T2.
No serious attempts were made to maximize performance.
Click on the image for a larger version.

Figure 1 (above) depicts the electrical diagram of the amplifier that was literally constructed on the workbench using a lot of clip leads and "floating" components as shown in the pictures.  Because this was a quick "lash-up" I used components that I had kicking around with no real attempt whatsoever to obtain maximum performance.

The audio source for this was my old NexBlack audio player, designed to drive only a standard pair of 32 ohm headphones.  To get some voltage gain and to obtain the 180 degree phase split to provide differential drive to the pair of tubes I fed the audio into one of T1's 5 volt secondaries with the grids connected to the dual 120 volt primaries in series, using the middle as the center-tap to which a "bias cell", a single 1.5 volt AAA cell, was connected to provide some negative voltage.

Even though T1 was a simple split-bobbin dual primary, dual secondary power transformer, it worked reasonably well in the role of audio transformer.  With the 5 volt to 240 volt secondary and primaries, the turns ratio was approximately 1:48 implying a possible impedance transformation of 2304-fold across the entire "secondary".  In this application the actual impedance is not important as it was only the "voltage gain" and the 180 degree phase split that was sought.  In the configuration depicted in the Figure 2 there was more than enough drive available from the audio player to drive the tubes' grids into both cut-off and saturation.

Both V1 and V2 were wired in "triode" configuration with the screen (G2) tied to the plate supply and the audio and operating bias being applied to the first grid.  Because these tubes' filament voltage is specified to be in the range of 0.9 to 1.2 volts, a 4.7 ohm series resistor, R1, was used to drop the filament voltage from NiMH cell B2 to a "safe" value of about a volt.  The plate voltage was provided by five 9-volt batteries in series with a bench supply to yield around 60 volts - the recommended voltage for this particular tube.
Figure 3:
The amplifier, wired up and scattered across the workbench.  The audio
player and T1 are along the left edge, the tubes are in the middle and
the output transformer, speaker and batteries that make up the
plate supply are seen to the right.
Click on the image for a larger version.

In the same spirit as T1 the output transformer was also one designed for AC mains use rather than an audio transformer.  In trying a number of different transformers that could be wired with a center-tap on the highest-voltage winding - including the same type as used for T1- I observed that the highest audio output power was obtained when I used the plate voltage transformer that I'd wound for a (yet to be described) audio amplifier that I'm constructing.  (For an article about the construction of this transformer follow this link).

For T2 this transformer was used "backwards" with the 982 (unloaded) volt center-tapped secondary being connected to the tubes' plates in push-pull configuration.  With a tone generator being used as the audio source I experimented with the various taps and winding combinations and found that the best speaker drive was obtained across the "ten volts" of the 115 and 125 volt taps of the primary.  Based on this configuration the calculated turns ratio is therefore around (982/10) = 98:1 implying an impedance transformation of 9604:1.  With the 8 ohm speaker, the total impedance across the entire winding is therefore calculated to be approximately 77k, or around 19k between the center-tap and each end.  In rummaging around I noted that this particular transformer appeared to have the largest turns ratio of any that I had on-hand!

Perhaps due to the "open" construction and flying leads and/or the lack of any swamping/terminating resistance on the grid side of T1 I noted on the oscilloscope some high-frequency oscillation on the audio output which was easily quashed with the addition of 100pF capacitors C1 and C2 on the grids of the tubes.  The addition of C3 as a power supply bypass had a very minor affect, slightly improving the amplifier performance as well - such as it was!
Figure 4:
A close up of the two tubes, flying leads, C1 and C2 and filament
battery B2 in the background.
Click on the image for a larger version.

In initial testing bias cell B1 was omitted resulting in a quiescent current of around 6 milliamps with 60 volts on the plates.  Adding this cell  to provide a bit of negative bias lowered this current to around 2.5 milliamps while also improving the output power capability somewhat.  Increasing this bias to about -3 volts (two cells in series) resulted in lower audio output and a noticeable amount of crossover distortion indicating that too much of each audio cycle was occurring where the tube's linearity suffered and/or it was in cut-off.

The audio output power was a whopping 250 milliwatts or so at 1 kHz and approximately 10% distortion while the saturated (clipping) output power was around 550 milliwatts.  Referenced to 1 kHz, the -3dB end-to-end frequency response was approximately 90Hz to 12kHz with a broad 3 dB peak around 6 kHz.  On the "full-range" 6"(15cm) speaker that was used for testing this amount of power was more than loud enough to be heard everywhere in the room and sounded quite good with both speech and music.  If I had used a higher-power "rod" tube like the 1J37B or 1P24B and adjusted the impedance accordingly I could have gotten significantly more output power from this circuit.

While the overall frequency response performance could have been improved somewhat with more appropriate termination of transformer T1, one cannot reasonably expect the use of transformers intended for 50/60 Hz mains frequencies to provide the the best frequency response and flatness - particularly with the high plate impedances of the output.  Having said this, it is worth noting that power transformers such as that used for T1 may not only be used as a driving transformer but it could have also been used as an output transformer in a push-pull configuration, albeit with a lower impedance and audio output power for these particular tubes.  While the performance may not be ideal, these power transformers worked surprisingly well and their price, variety and availability make them suitable candidates for a wide variety of applications!

After satisfying my immediate curiosity about these tubes for the moment I un-clipped the flying leads, unsoldered the capacitors and resistors and put the parts away.  Some time in the future I'll put together a few more "fun" projects using these interesting tubes.

[End]

Monday, December 19, 2016

On the winding of power chokes and transformers: Part 3 - The plate (high voltage) transformer

This is a follow-up of two previous posts in this series:
  • On the winding of power chokes and transformers: Part 1 - Chokes - link
  • On the winding of power chokes and transformers: Part 2 - A filament transformer- link

Using what we already know:

Figure 1:
Plate transformer with attached wires and end bells installed.
The windings and laminations are yet to be varnished or the end bells painted.
Click on the image for a larger version.
In the previous post of this series I described the design and construction of a filament transformer with dual 11 volt, 11 amp windings and a multi-tapped primary.  Building on the experience gained, I felt confident to take it to the next step:  The design and building of the high voltage "plate" transformer for the (yet to be described) tube amplifier.

Based on the characteristics of the tubes to be used, the plate voltage needed to be "around 1 kilovolt" with each amplifier section requiring "about 100 milliamps" of average current, or around 200 milliamps, for the pair of channels.  Because of the experience gained in the winding of the filament transformer, we could use the design of the primary winding as a starting point.  For example, we know that to achieve a target magnetic flux of 1.4 Tesla and have the transformer be capable of at least 253 volt-amps and attain a rather conservative cross-sectional amperage of 0.4 amps/mm2 for the primary winding's we could use:
  • 17 AWG wire
  • Taps at 220, 229 and 239 turns for 115, 120 and 125 volts respectively, at 60 Hz
During the winding of the filament transformer it was observed that I could easily fit 41 turns of 17 AWG per layer.  This meant that the 239 turns only partially filled the final (fifth) layer, so we could afford to add a few more turns to the primary if necessary.

Two secondaries needed:

While the main secondary will be for high voltage, we will also need a 6.3 volt secondary to power the filaments of some of the driver tubes.  Because such a secondary will have relatively few turns we will need to calculate it first, for reasons that will become clear.

Using the "5% rule" we calculate that our 6.3 volt secondary will actually need to produce 105% of the desired voltage (6.3 * 1.05) = 6.6 volts to account for the drop under load.  Taking our 229 turn, 120 volt primary as a starting point we determine that the turns ratio to achieve this voltage would be (120 / 6.6) = 18.182:1 turns ratio.  With our 229 turn, 120 volt tap we would need (229 / 18.182) =  12.59 turns to obtain 6.6 volts.  

What this means is that for our secondary we should round the number of turns up (I'll explain why shortly) rather than down and with exactly 13 turns we end up with a primary-secondary turns ratio of (229 / 13) = 17.62:1.  From this we can calculate the actual, unloaded secondary voltage will be (120 / 17.62) = 6.81 volts - a bit higher than we'd like.

How do we fix this?  We should increase the number of turns on the primary to be able to more accurately obtain the desired voltage, by why increase the number of turns when we could also establish an accurate result by rounding down the secondary to, say, 12 turns and decreasing the number of turns on the primary to compensate?

You may recall that when winding a primary, the magnetic flux is has an inverse relationship with the number of turns.  Because the number of turns on the primary of the filament transformer was calculated to achieve the maximum target flux, we would not want to decrease the number of primary winding turns as that would increase that flux.  In other words, the main down side of adding a few turns to the primary is that each winding will need a proportional number of extra turns as well, taking up additional room on the bobbin:  If things are already tight, adding those turns could result in more wire than will fit.

Crunching the numbers:
  • Our voltage ratio:  120 / 6.6 = 18.182:1.  We already saw this number.
  • Since our 6.6 volt secondary should have exactly 13 turns, our 120 volt primary should have (18.182 * 13) =  236.4 turns, rounded down to 236.  This increase in turns reduces the magnetic flux from 1.4 to about 1.3 Tesla.
Clearly, a half a turn on the 120 volt winding has a fraction of the effect (18.182th, to be more precise) as a half turn on the low-voltage primary so we will round this down to 236 turns.  Let us now calculate the 115 and 125 volt taps:
  • 115 volts / 6.6 volts =  17.42:1 ratio.  13 turns * 17.42 = 226.46 turns.  I rounded this down to 226 turns.
  • 125 volts / 6.6 volts = 18.94:1 ratio.  13 turns * 18.94 = 246.22 turns.  This was rounded down to 246 turns.
Since we already know from when we wound the filament transformer that we can safely put 41 turns on a layer, we can see that for 246 turns we would need (246 / 41) = 6.0 layers - so we will go with that!

Designing the high voltage secondary:

If you are familiar with tube-type amplifiers you may have already have guessed from the voltage and current requirements that the plate impedance of the amplifier would be quite high:  10k ohms, to be precise.  The output transformers themselves are designed for single-ended triode operation with 8 ohm secondaries, rated for 25 watts (maximum) output.  Going through the math one can see that the turns ratio of this transformer is approximately √(10000/8) = 35.36:1.  If 25 watts RMS were being produced into 8 ohms, this implies that the RMS output voltage is around 14.14 volts, or almost exactly 500 volts RMS on the 10k primary which translates to 707 volts peak.

According to the specifications gleaned from the Edcor support forum (a link to the message thread may be found here) the maximum "safe" voltage across the primary and secondary windings would be 1000 volts.  Clearly, assuming a 10k primary impedance, 25 watts RMS of power and any reasonable plate voltage to achieve anywhere near this output power one will have to exceed this maximum voltage rating - unless a bipolar power supply is used where the high voltage is split - that is, the standing DC voltage between the primary and secondary is reduced to half.  To do this a full wave "bridge" rectifier is used with our choke-input filter network with the centertap of the transformer being grounded.

A final (loaded) DC voltage of around 970 volts for the plate voltage was (somewhat arbitrarily) decided as the target for the tubes that will be used - a reasonable compromise between the constraints of the output audio transformer voltage rating and the efficiency of the tube.  With this in mind let us calculate the actual, unloaded voltage for the secondary.

We know from when we designed our choke that at 200 mA there will be a 60 volt drop, so we will need to increase the output of 970 volts by this amount, which means that we will need (970 + 60) =  1030 volts.  Because the power supply will use a choke input we know that the loaded voltage of such a power supply is typically around 110% of the RMS voltage which means that for 1030 volts DC we will need approximately (1030 / 1.1) =  936 volts RMS.

Using the "5%" rule of thumb to take into account resistive loading of the primary itself we can calculate the actual, loaded voltage for the secondary, as in (936 * 1.05) =  982 volts, unloaded.  Using the 120 volt tap from the reference design we can now calculate our turns ratio and the number of turns, as in:
  • 982 volts / 120 volts = An 8.183:1 turns ratio.
  • 236 turns (at 120 volts) * 8.183 = 1931 turns which will be rounded down to an even 1930 turns so that the center-tap will be made at the 965th turn.
Based on the recommendations from the Turner Audio and Homo-Ludens web pages (see previous articles for the links) we can use a general rule of thumb of 0.33-0.35mm2/amp and since our current is to be 0.2 amps, we need a wire with the size of at least (0.2 amps * 0.33 mm2/amp) = 0.066 mm2.  Consulting our wire chart we see that 29 AWG has a cross-sectional area of 0.0642 mm2 resulting in a density of 0.321 mm2/amp - pretty close to our design goal.  As noted in the previous installment, Edcor seems to use a value of around 0.253 mm2/amp for their transformers and if this is applied our primary would be capable of (0.0642 mm2 / 0.253 mm2/amp) = 0.25 amps.

As it happens I had 29 AWG wire available when the choke was wound (it, too, was designed for 200mA) so this is the wire that I used.

Will it fit? 

At this point the question must be asked:  Will all of these windings fit on the bobbin?

We know from when we wound the choke that approximately 161 turns of 29 AWG wire will fit per layer, and with 1930 turns total, we'll need 12 layers.  With 29 AWG wire having an outside diameter (with insulation) of 0.33mm and the tape from each layer adding 0.05mm of thickness, each layer will occupy 0.38mm or, with 12 layers, 4.56mm of of bobbin "height". 

We also know from our winding of the filament transformer that one layer of 17 AWG wire plus 0.05mm of insulating tape has a total height of 1.274mm and with 6 layers (yes, I know that there is actually five full layers) that comes to 7.644mm.  Put together, the combined height of both sets of windings is 12.204mm - approximately 73% of the 16.5mm available bobbin height.


Figure 2:
Center tap of high voltage plate winding located in the middle of the winding
before Nomex insulation was added.
Click on the image for a larger version.
This figure does not include the low voltage secondary winding (one layer of 17 AWG, adding another 1.274mm) or the extra insulation that must be added between windings (approximately 0.5mm for each of the three) all of which adds another 2.774mm, taking us up to 13.704mm - about 83% of the available space.

While this will be kind of a tight fit, we ended up with the same sort of numbers when we built designed and successfully built the filament transformer so we can have good confidence that this, too, will work.

The winding:

While it may seem customary to wind the primary first, that may be because most transformers that are seen these days are step-down, with the secondary winding handling more current than the primary and thus using larger wire.  It usual to place the smallest wire on the inner-most winding since it is more flexible and  easier to handle on the smaller-diameter "inner" layers of a bobbin, going around the square-ish corners and leaving the larger wire for later when the bobbin diameter is larger and the corners more rounded.

Following this convention a hole was "drilled" in the side of the nylon bobbin with a hot soldering iron and a piece of Teflon™ insulated wire was pulled through, attached to the start of the winding and then insulated with several layers of polyimide tape and a layer of Nomex™ paper insulation.  With that task completed the winding proceeded carefully with care being taken on the first layer to assure both neatness and tight packing - the latter being done by pausing every few turns to slide the wire over to minimize the gap between adjacent conductors.

Figure 3:
End of the high voltage secondary winding, insulated with both
polyimide tape and Nomex ™ paper.  A loop was made in the wire which
brought out at a right angle from the other turns so that the tap would
not interrupt the continued neat, side-by-side windings.  Taps are
always made on the two sides of the bobbin that face the end bells
rather than the sides inside the core where the height is
is more limited.
Click on the image for a larger version.
The first layer done, a single layer of 0.05mm polyimide tape was placed over the top.  When I wound the choke I had only a single width of this tape available, but this time I had a selection of widths so as I proceeded with the layers, the location of the overlap and widths of this tape was changed with each layer to minimize "piling" of the turns where the tape overlaps which would later make it difficult to keep the layers even.

After a few hours of intermittent winding over several days - with each layer individually insulated with 0.05mm polyimide tape - the center tap was reached and for this a loop of wire was made in the conductor at right angles to the lay to which another piece of Teflon wire was soldered which was brought through the side via a hole made in the side of the bobbin with a hot soldering iron.  This joint was carefully placed in the middle of the flat side of the bobbin that would face outward from the core and insulated it with a few layers of polyimide insulation and Nomex paper to prevent it from damaging or being damaged by the pressure of turns in the layers above and below.
Figure 4:
Overlay of Nomex ™ insulating paper atop the finished high

voltage secondary winding before the top layer of polyimide
tape and its "creepage" insulation along
the sides of the bobin was added.
Click on the image for a larger version.



After a few more days of occasional winding the last turn was laid down, nearly filling the 13th and final layer.  I soldered to this a piece of Teflon wire and insulated it and the wire was brought out through the side of the bobbin and the entire secondary was covered with several layers of polyimide tape and 0.05mm Nomex paper.  As a final covering over the Nomex, another layer of polyimide tape was laid down, this time with the tape slightly going up the sides to increase the "creepage" distance between the primary and secondary - a sensible safety precaution, particularly with a high-voltage transformer!

Now, the primary...

The conductors of the primary were now laid down atop the insulated secondary.  As with the filament transformer the 17 AWG wire was brought directly out through the side of the bobbin and tucked out of the way:  The connection to flexible wire would be done later.
Figure 5:
The three "end" taps of the primary winding:  Top-left is the 115 volt tap,
below it is the 120 volt tap with the 125 volt finish on the left.  After
this picture was taken small pieces of Nomex paper and additional
tape were placed below and above the taps.
Click on the image for a larger version.

As with the start of any new winding the first layer of the 17 AWG primary was done with special care to make it neat and tight and each layer was individually insulated with 0.05mm polyimide tape.  When the 220th and 229th turns (for the 115 and 120 volt taps, respctively) were reached, loops of wire were put in the conductor, which was brought out through marked holes in the bobbin at right angles to the conductor.

With each tap being insulated with polyimide tape and Nomex paper where they crossed over other windings, the entire primary was then covered with several layers of polyimide tape and Nomex paper.  Again, a bit of insulation was brought up along the sides of the bobbin to provide extra "creepage" distance to provide good insulation for the 6.3 volt secondary to maximize both safety and reliability.

More about the 6.3 volt secondary winding:

Because it was on-hand, 17 AWG wire was used for the "6.3 volt" additional secondary.  With a cross-sectional area of 1.04mm2, we can calculate its current-handling ability:
  • Using the 0.33 amps/mm2 recommendation from the Turner Audio site, a safe current is:  (1.04mm2 / 0.33 amps/mm2) = 3.15 amps
  • Using the 0.253 amps/mm2 design Edcor guidelines a safe current is:  (1.04mm2 / 0.253 amps/mm2) =  4.11 amps.
Figure 6:
The completed winding - including the 13 turn, low-voltage secondary -
with the just-started core stacking.
Click on the image for a larger version.
Even in the worst-case scenario the addition of a 4.11 amp secondary would add only another 28 volt-amps of load to the transformer - well within its capacity.  Because this winding is on the outside of the bobbin and "exposed", it has good opportunity for cooling by convection and thus the Edcor rating would seem to be applicable - and 4 amps is plenty of current for several 6.3 volt tubes.

Comment:  If more current is needed it would be easy to add another parallel 17 AWG conductor to double its capacity.

As with the primary winding - which also used the same 17 AWG conductor - the ends of this 13 turn secondary were brought straight out the sides of the Nylon bobbin for later connection to flexible conductors and this additional secondary was overcoated with polyimide and polyester tape.

Finishing and initial testing:


With the addition of the low voltage secondary, all layers were over-wrapped with another layer of polyester tape to both secure and insulate the windings.  The transformer was almost ready to be tested!

Figure 7:
The stacked transformer undergoing initial testing with a
a variable transformer.
Click on the image for a larger version.
Although there are approximately 111 pieces of iron to be inserted into the core, the process is pretty easy:  Simply lay the bobbin on the table on one of the "outer" faces (where the taps are made and wires are attached) and alternately place the "E" sections atop each other.  With the "E" sections done, the transformer is then set on end to provide access to the vacant slots between every other "E" section into which the "I" sections were dropped.  Once these sections were added to one side, the bolts were slid through the laminations with the "I" sections to prevent them from falling out as I turned the transformer over and the final pieces were added to the other side.

With all E and I sections installed, a block of wood and a small hammer were used to abut the pieces of laminations against each other, a process that required several passes on all four sides.  With this done some nylon shoulder washers were installed (visible under the screw heads in Figure 7) to prevent the effect of eddy currents that might be caused by the "shorted turn" effect of the screw, and the bolts tightened.

Using a variable transformer the transformer was then tested, first noting that the unloaded (magnetization) current of the transformer was comparable to that of the previously-tested filament transformer indicating that there seemed to be nothing amiss.  Very carefully, the high voltage secondary's voltage was then tested on each side of center tap and I noted that they were within a fraction of a volt of each other, and exactly at the calculated value with 120.0 volts applied:  491 volts on average.  I could not directly measure the 982 (unloaded) volts across the entire secondary since I have no voltmeter that is "officially" rated above 750 VAC.

After a test of the low voltage secondary, which was also measured to be at its designed voltage, I attached permanent wires and the end bells as seen in Figure 1 at the top of this page.  At this point the transformer  only awaits being dipped in insulating varnish - something that will happen after inital testing of the (yet to be described) amplifier prototype.


A future post in this series will describe the final steps in finishing these transformers:  Impregnation in "insulating varnish" and the final painting of the end bells.

[End]

Friday, December 9, 2016

Repairing the power switch on the Kenwood KA-8011 (a.k.a. KA-801) amplifier

Back around 1990 my brother mentioned to me that there was an amplifier, in a box, in pieces, in the back room of the home TV/electronics store where he worked at the time and that if I made an offer I could probably get it for cheap.  Dropping by one day I saw that it was a Kenwood KA-8011 Integrated DC amplifier (apparently the same as the KA-801, except with a dark, gray front panel) laying in a box from which the covers were removed with a bunch of screws and knobs laying in the bottom.  I also noticed with some surprise that it had a world-wide voltage selector switch on the back and that the power cord had a Japanese 2-prong wall plug and U.S. adapter - and still does!  All of the parts seemed to be there so I offered some cash ($50, I seem to recall) and walked out with it and a receipt.
Figure 1:
 Spoiler alert:  This is the KA-8011 with the repaired power switch.
As noted in the text, the original, blue-painted panel meter lights were
replaced long ago with blue LEDs.


When I got home with the amplifier I knew that I had my work cut out for me particularly since, in those days, before the widespread internet, I had no schematic for it and no-one that I contacted seemed to be able to find one.  Powering it up I noted that the speaker protection relay would never engage indicating that there was a fault somewhere in the amplifier.

A visual inspection of the awkward-to-reach back panel's circuit board revealed several burned-looking leads sticking up from the circuit board where transistors had exploded and several burned resistors.  After a few hours of reverse-engineering a portion of the circuit I realized that the majority of the circuit at fault was one of four identical phono preamp input circuits (there are two separate stereo phono inputs) and associated low-level power supplies.  Between the intact amplifier sections and being able to divine the color bands on the smoked resistors - along with some educated guesses - I was able to determine the various components' values and effect a repair.
Figure 2:
The power switch, with a broken bat.
Click on the image for a larger version.

The amplifier now worked... sort of.  I then had to sort out a problem with the rear-panel input selector switch, operated by a flat, thin ribbon of stainless steel in a plastic jacket that was engaged from a front-panel selector.  I managed to cut off the portion at the front that had been damaged where it was pulled-on from the front panel having been loose in the box, punch some new holes in the ribbon, align the two (front and rear) portions of the switch mechanism and restore its operation.

Snap!

Having done the above, the amplifier was again operational and I have used it almost every day in the 25 or so years since, needing only to replace the blue-colored incandescent meter lights with LEDs, powered from a simple DC filtered supply.  In the intervening years I also had to replace some of the smaller electrolytics on the main board that had gone bad, causing the speaker protection circuit to randomly trip on bassy audio content and with slight AC mains voltage fluctuations.

Figure 3:
Comparing the old (top) and new (bottom) switch components.  In order
to prevent it from interfering with the body of the switch some of the
metal on the new bat would have to be removed.
Click on the image for a larger version.
I was annoyed when one day, a few months ago, the power switch handle - which had been bent before I got the amplifier - and then "un-bent" during the repair - broke off in my hand when I turned it on.

Using the "bloody stump" of the power switch for a few months  I finally did a search on EvilBay to look for a new switch.  While I didn't find a power switch I did see a "tone control" switch for the same series of amplifier - so I got that, instead.  When it arrived I noted, as expected, that most of it did not mechanically resemble the power switch or look as though it would easily mount in the same location, but it did have essentially the same metal bat on the end as the original that I figured I could fit onto missing portion that had broken off the power switch.

Comment:
Even though the "new" switch was much too small - of insufficient current rating - to have been used to switch the mains (AC input) power, it would have sufficed to operate a relay.  To have done this would have required that new holes be drilled in the front sub-panel to match those of this new, smaller switch. While this would not be "original" circuitry, it would have looked the same from the front panel and is a possible option should the power switch itself become unreliable some time in the future.

Removing the original power switch I laid the two side by side and made notes of the differences between and the metal bat of the original, which was narrower in some places to clear parts of the switch body, and taking a file to the new one I took off some metal to clear the possible obstructions.  I then noted on a crude drawing the length and orientation of the new bat based on the axis of the switch's pivot point.  Because the bat of the original switch was embedded in a block of molded Bakelite I knew that I would have to somehow attach a portion of the new switches' bat to the old, so I carefully disassembled with old power switch, cutting off and saving the original rivet on which the switch pivoted, and carefully noting where everything had gone, saving the small springs, contacts and some small Bakelite pins.
Figure 4:
The new bat, butt-soldered on the old switch.  Note that the bat from the
"new" switch has been filed to better-resemble the shape of the
original bat to clear the switch body.
Had I not been able to find a "similar" switch on EvilBay I could have
probably measured the original switch, found some scrap
steel of similar thickness and made a suitable replacement entirely
by hand with careful filing using another switch as a template.
Click on the image for a larger version.

Clamping the old part in a vise I cut off most of the original bat, leaving about 5mm of metal remaining.   Carefully comparing the old and new piece I then marked where, on the new bat, that I would have to cut to allow the repaired piece - consisting of the new and old butted and laid end-to-end - have the same length as the intact original.  Doing so - purposely cutting the "new" bat slightly long - I did some fine tuning with a file until the two pieces laid down precisely lined up as they should.

Attaching the new piece

Using some silver solder intended for stainless steel I applied some of its liquid flux - apparently a mixture of chloric and hydrochloric acid - and using a very hot soldering iron, "butt-soldered" the two pieces together in careful alignment and then filed the surfaces flat to remove excess.  While the bakelite switch body can handle a brief application of a soldering iron, I knew that it would not tolerate the heat from a proper, brazed joint.

This (weak!) solder joint was intended to be temporary, need only to be good enough to allow a sleeve to be made by wrapping an appropriately cut piece of thin, tin-plated steel (from my junkbox) around the joint.  Once this sleeve was checked for proper fit and folded tightly, additional flux was applied and the entire joint - sleeve and all - was soldered, the result being a very strong repair with the restored bat being of the same length and at the same angle as the original.

Reassembly:

The trick was now to get every thing back together.

Figure 5:
The steel sleeve being installed over the butt solder joint,
before final folding and soldering.
Click on the image for a larger version.
Reinstalling the pivot and making a few clearance adjustments to the original switch's frame with a small needle file, the original rivet was then soldered into place and the entire assembly washed in an ultrasonic cleaner to remove the remnants of the corrosive flux from the bat and switch body.

In the base of the switch, the contacts, which were the same as those had it been an SPDT switch, were reinstalled - this time, rotated 180 degrees so that the previously unused contact portions would now be subject to electrical wear.  These contact were then "stuck" into place with a dab of dielectric grease so that they would not fall out when the switch body was inverted.

Figure 6:
The repaired switch, reassembled,  with the new bat spliced on.
Click on the image for a larger version.
After reinstalling the springs and pins, the rear part of the switch with the contacts was placed over the top of the moveable portion, held in the mechanical center, and the base was carefully pushed into place, compressing the internal springs and pins.  Holding everything together with one hand the proper operation of the switch was mechanically and electrically verified before bending the tabs to hold everything into place.

In reality the reassembly didn't go quite as smoothly as the above.  During one of the multiple attempts to get everything back together the smaller-diameter rear portion of the small, spring-loaded Bakelite pins used to push on the contacts snapped off.  To repair these pins the front, larger-diameter portions - that which pushed against the metal contacts - were placed in the collet of a rotary tool and a shallow hole was drilled into the rear portion where the broken pieces had attached to fit short pieces of 18 AWG wire:  By rotating the piece into which the hole was to be drilled, the exact center is automatically located.  The pieces of wire were then secured using a small amount of epoxy - a process accelerated by placing the pins in a 180F (80C) oven for an hour.  After the epoxy had set the wires were then trimmed to the length of the original sections that had broken off and the ends smoothed over with a small needle file to prevent their snagging on the spring.  The result was a repair that was stronger than the original pins and these easily survived the reassembly.

The results:

Figure 7:
After reassembly it was noted that the gray "skirt"
was hitting the front sub-panel frame, preventing it from
being set to the "off" position.  A bit of heat was applied to
set a permanent bend so that it would clear this panel.
Click on the image for a larger version.
The amplifier was then put back together, very carefully.  The only real issue that I noted was that the gray plastic skirt/escutcheon on the bat ended up about half a millimeter farther away from the switch body and closer to the sub panel than before, causing it to snag on the front sub-panel's cut-out when I attempted to move it to the "off" position.  Careful softening of the plastic with the rising heat of a soldering iron and bending it very slightly allowed it to clear.

Putting all of the knobs back on, tightening the bushing nuts and screws as necessary before doing so, I then tested the amplifier on the bench and was pleased to find that I'd not managed to break anything.

Finding that everything was working fine I put it back on the shelf where it belongs where I continue to use it often.

[End]

Monday, November 28, 2016

On the winding of power chokes and transformers: Part 2 - A filament transformer

Having wound the choke described in the previous installment about chokes - (link) - I decided to proceed with the next logical step in the project:  Winding a filament transformer.  Also see the follow-on article Part 3 - The Plate (High Voltage) Transformer - (link).

With the lower voltage requirements, the filament transformer is the next-easiest since being a step down transformer, fewer turns are required overall and the wire sizes will be larger.

The first step was to figure out my voltage and current requirements - but this was already known in the form of the filament requirements of the tubes to be used:  Two center-tapped windings, each capable of 11 volts at 11 amps.  To calculate the necessary winding parameters (e.g. number of turns, size of wire, etc.) I will refer again to the two links noted in the previous installment, included below:

  1. Turner Audio (link) - These pages contain much practical advice on power and audio transformers and chokes.  (Refer to the link "Power Transformers and Chokes" (link) and related pages linked from that page.)
  2. Figure 1:
    The completed filament transformer, before varnishing,
    ready for testing.
    Click on the image for a larger version.
  3. Homo-Ludens - Practical transformer winding (link) - While mostly about power transformers, this page also contain practical advice based on hands-on experience of winding, re-winding and reverse-engineering/rebuilding transformers.  There is also another linked page "Transformers and Coils" (link) that has additional information on this topic.
While there are enough equations and general information spread across both pages to provide the necessary information if you want to crunch numbers with equations, of particular interest is a spreadsheet found on the Homo-Ludens "Practical transformer winding" web page that allows one to "play" with various configurations.  For this spreadsheet we will need to input what we already know, such as:
  • Input voltage:  120 VAC (nominal) at 60 Hz.  Since we want to have multiple taps to fine-tune the voltage, we'll also calculate for 115 and 125 volts.
  • Output voltage:  11 volts under load.  A rule of thumb is to add 5% to this to accommodate various losses so this would be (11 * 1.05 = 11.025) or approximately 11.5 volts.
  • Output current:  22 amps - the total sum of the two 11 amp filament windings.  They will be "split" in later calculations.
  • Core size:  E150.  The Edcor core and bobbin that will be used has a stack height of 38mm and a center leg that is 38mm across.
  • Set a design goal for wire sizes corresponding with a current density of 0.4 mm2/amp, a rather conservative number.
  • Let us initially set a "fill factor" of 0.4 - more on this parameter, later.
  • Core material information:  The Edcor laminations use M-6 GOSS (Grain-Oriented Silicon Steel) which is a material that is capable of safely handling higher magnetic flux than "generic" iron cores.  This has two important implications:
    • The saturation flux of this material is in the area of 1.7 Tesla.  This is a very "soft" number, dependent largely on how much core heating one is able to tolerate in the intended application.
    • The iron loss (in watts/kg at 1 Tesla) for the M-6 material is quite low - approximately 0.5 watts/kg@1T (at 50 Hz) versus 2 watts/kg@1T for "generic" transform iron.  The spreadsheet expects a 50 Hz value here regardless of the actual frequency.
A few words about the wire size:

The value of 0.4mm2/amp target that I chose is fairly conservative based on the recommendations found in several sources:
  • The Turner Audio site suggests a value of (3 amps/mm2) = 0.33mm2/amp as a general number.
  • The Homo-Ludens site suggests a value of 0.35mm2/amp for "medium-sized" transformers (50-300 watts) wire such as this and smaller/heavier (0.25 and 0.5mm2/amp) conductors for very small and large transformers, respectively.
  • Various vintages of the ARRL Amateur Radio Handbook note that a value of 1000 cma (0.506mm2/amp) as being "conservative" with a value of 700 cma (0.354mm2/amp) being suggested.
  • Interestingly, the 1936 Jones Radio Handbook notes recommends a 1000 cma
    (0.506mm2/amp) value for typical amateur use and increasing this to 1500 cma (0.759mm2/amp) for transformers that would be intermittently subjected to significant overload and/or were in hot, poorly ventilated environments.  These recommendations are understandably based on the use of older materials such as paper insulation and the more fragile varnished/enameled wire of the day.
  • If one peruses the Edcor site one can glean bits of data here and there and they mention a design goal of 500 cma (circular-mill amperes) which converts to 0.253mm2/amp.  (Reference:  Tek Note 43 - link.)  When I read this I presumed that this recommendation may have been intended for small, low-power transformers, but I noted this posting - link in their forum where a current of 200mA is mentioned being used with 30 AWG wire which calculates to 0.254mm2/amp.
Even more about flux density:
  • As noted, for inexpensive, generic cores of unknown properties Turner Audio suggests a maximum flux of 0.9 Telsa while the Homo-Ludens site suggests that 1.0 Tesla is "probably OK" for the vast majority of cores of unknown provenance.  The later site recommends that if these cores are being re-used that one counts the number of turns on the original primary (if it is being re-wound) and use this, along with the core's cross-sectional size to estimate the original flux density.
  • The M-6 material is capable of much better performance (e.g. lower loss) than "generic" iron - likely being usable at 1.6-1.7 Tesla, but Edcor mentions in Tek Note 43 (linked above) that their design goal is 1.4 Tesla - value with which both the Turner Audio and Homo-Ludens sites agree as being appropriate for this material.  Based on typical curves for M-6 material, this would seem to be a reasonable compromise between higher core losses, fewer turns (e.g. higher flux) and more turns with higher copper losses, lower core losses (lower flux).
 Based on the above I decided to use 1.4 Tesla in my design.

Comment:

If you are keeping the original primary winding, wind a few dozen turns of hookup wire and carefully measure the resulting, unloaded voltage.  Comparing this with the applied primary voltage and taking the number of temporary turns that were wound the number of turns on the primary could be quite accurately determined and from there, along with the stack height and center leg size, it should be possible calculate the approximate magnetic flux of the original device.
Crunching the numbers:

Inputting the above to the spreadsheet one can see that it does not actually care about the output current, but rather is tells you the highest possible load current and volt-amp capacity based on the core size and flux density that you specify and the most important information that it gives is the number of turns for the primary:  It is up to you to scale back the "worst case" numbers that it gives you to better suit your needs and make sure that everything will fit in the available space.

For example, given the information that we already have, the spreadsheet calculates that with the entered parameters one could expect to pull well over 26 amps at 11.5 watts - about 292 volt-amps using the wire targets along with what is calculated to be able to fit given the calculated wire sizes and the inputted fill factor.  In reality, we will need closer to (11.5 volt * 22 amps =) 253 volt-amps so we would be safe in downsizing our wire to about 83% of the calculated cross-sectional area.  Assuming the worst case loading of the primary - which occurs at the lowest primary voltage, 115 VAC, we can calculate that our maximum primary current will be (253 volt-amps / 115 volts) = 2.2 amps.
  • If we consult a wire table to see which size most closely matches our 0.4mm2/amp criteria (e.g. 0.4mm2/amp * 2.2 amps = 0.84mm2) we find:
    • 17 AWG wire at 1.04mm2.  This is (1.04mm2 /amp / 2.2 amps) = 0.472 mm2.
    • 18 AWG wire at 0.823mm2.  This is (0.823mm2/amp / 2.2 amps) = 0.37 mm2.
    • 19 AWG wire at 0.653mm2.  This is (0.653mm2/amp / 2.2 amps) = 0.30 mm2.
As we can see, either 17 or 18 AWG would be fine for the primary, both sizes being quite close to our design goal:  17 AWG will run a bit cooler with lower loss while 18 AWG will take up a bit less space on the bobbin.  19 AWG does fit within the Edcor guidelines but is much smaller than target - but would still probably be OK if one tolerated a bit of extra heat and voltage drop.

Based on the 1.4 Tesla flux values we can see that at 115 Volts we would need 223 turns on our primary to achieve the target of 11.5 volts and since the ratio of primary-secondary turns is exactly the same as our voltage ratio, we can calculate:
  • 115 volts / 11.5 volts = 10:1 ratio
What this means is that for our 223 turns on the 11.5 volt primary, we would need (223 / 10) = 22.3 turns.  Since it is awkward to wind a fractional turn, let's round down to 22 turns - an even number that also makes it easy to locate the center tap point.  By increasing the number of turns slightly we must now recalculate the 115 volt primary winding using the same ratio as above:
  • Doing this, we will need (10 * 22) = 220 turns.  This reduction in turns from 223 increases the flux density on the core, but only by a few percent so we can ignore it.
Let us now calculate the number of turns for 120 and 125 volts:
  • 120 volts / 11.5 volts = 10.435:1 ratio.  22 turns * 10.435 = 229 turns, rounded down.
  • 125 volts / 11.5 volts = 10.870:1 ratio.  22 turns * 10.870 = 239 turns, rounded down.
Since we need two filament windings, each capable of of 11 amps, we calculate the appropriate wire size for each:
  • For 11 amps, we calculated a minimum wire cross-sectional area of (0.4 mm2/amp * 11 amps) = 4.4 mm2.  Consulting the table, we find:
    • 10 AWG wire at 5.26mm2.  This is (5.26mm2/amp / 11 amps) = 0.48 mm2
    • 11 AWG wire at 4.17mm2.  This is (4.17mm2/amp / 11 amps) = 0.38 mm2
    • 12 AWG wire at 3.31mm2.  This is (3.31mm2/amp / 11 amps) = 0.30 mm2
    • 13 AWG wire at 2.62mm2.  This is (2.62mm2/amp / 11 amps) = 0.23 mm2
From all of the above we can see the 11 AWG wire is very close to our 0.4mm2/amp target - and still above the recommendations of the two web sites listed above while 12 AWG appears to be suitable if one goes with the Edcor guidelines.  It should also be noted that because these primary windings are on the "outside" layer (the reason to be noted later) they can more readily dissipate heat via convection than a winding deep inside the bobbin.

Comment: 
Instead of using 11 AWG, I could have used four parallel strands of 17 AWG as they would have a total of (1.04 * 4) = 4.16mm2 cross-sectional area - although handling multiple conductors at once can be quite awkward.  One might do this if larger wire was not on-hand, but also to take advantage of the fact that 17 AWG is more flexible than 11 AWG.  When paralleling conductors care must be taken to make sure that all are wound identically to prevent the differences in their intercepted magnetic fields which can cause "bucking", resulting in heating.

Will it fit?

As it turned out I had suitably large quantities of 10, 11 and 17 AWG on hand so I decided to calculate the volume that would be taken up by the three sets of windings.  Based on online drawings of the Edcor E150 nylon bobbin - and actual measurements with a set of calipers - I came up with the following:
  • According to the drawing the interior width is 53.28mm but the actual, measured size was 52.7mm.
  • The indicated window "height" (e.g. the available space on one of the four sides into which the windings must fit) is 16.935mm, but the actual, measured size was 16.5mm.
First we calculate how many turns of 17 AWG will fit on a layer.  The wire that I used (polyimide coating, rated for operation to 200C) has a diameter with insulation of 1.224mm which means that (52mm / 1.224mm/turn) = 43.05 turns may fit in a layer.  Rounding down and accounting for about 1 turn of "fudge factor" (e.g. wire laying with a slight amount of space between adjacent turns, a slight bit of wastage at the ends where the next layer starts) we can reasonably expect 41-42 turns per layer.

Knowing that we will need 239 turns for the 125 volt winding this comes out to (239 turns / 42 turns/layer) = 5.7 layers so there should be no problem keeping it down to just 6 layers with a little bit of room to spare. Between layers I was laying down one layer of 0.05mm polyimide (Kapton (tm)) tape which means that for each layer I was taking up (1.224mm (wire) + 0.05mm (insulation)) = 1.274mm, and for 6 layers the total would be 7.644mm.  Between the primary and secondary we need to put at least 0.5mm of additional insulation, bringing that up to a total of around 8.144mm of height out of the available 16mm.

Now taking the 11 AWG secondary we note that the diameter of the wire with insulation is 2.393mm which means that (52mm / 2.393mm/turn) = 21.99 turns will fit on a single layer - and this number is a bit "soft" in that we may be able to squeeze the full 22nd turn in if the nylon bobbin will flex just a little. Using the above numbers we can see that each layer will take (2.393mm (wire) + 0.5mm (insulation)) = 2.893 mm - and since we have two identical windings that turns out to be 5.786mm, total.

All together, including a final 0.5mm thick layer of insulation, the height of the windings will be 13.93mm - about 84% of the available space and based on this I decided not to try the equations for 10 AWG. Out of curiosity I recalculated the above for 12 AWG we get (52mm / 2.139mm/turn) = 24.31 turns fitting on a single layer with each layer+insulation being 13.442mm - about 81% so this would have been fine but because since I had 11 AWG on hand I decided to proceed with that size.

It was noted in the aforementioned Edcor Tek Note 43 that a reasonable design goal is around a 70% filling of the bobbin but that at 90% the numbers are re-crunched to see if smaller wire may be used and/or a larger core is required:  Both of our numbers, above, come in below that 90% margin so we should be pretty safe if we are neat and careful.

Calculating winding volume using "Fill factor":

"Fill factor" is the ratio between the volume occupied by the wire itself and the combined volume of the wires and insulation.  Because a circle that is 1mm diameter occupies about 79% of the volume of a square that is 1mm on a side we lose over 20% off the bat in our packing efficiency - and this is made only worse by the fact that we need to add insulation between layers and also that we cannot pack the wires perfectly side-by-side.  A bit less easy to calculate is the fact that at the ends of the bobbin where we transition layers, we tend to lose a portion of each turn at each end.

On the Turner Audio pages it was noted that a "Fill factor" of around 0.3 was common with older transformers with (thick!) paper insulation between each winding and closer to 0.45 with modern insulation was practical while the Homo-Ludens site mentions that a fill factor of around 0.5 is practical if it is wound with care (e.g. neat, side-by-side windings) and one uses thin, modern insulation.

How does our transformer "stack up" when using this method?

We know from above that the window size is (52.705mm * 16.51mm) = 870mm2, so let us calculate how much of the bobbin our wire is expected to take up:
  • 17 AWG is 1.224mm diameter so its cross-sectional area is 1.177mm2, so (1.177mm2/turn * 249 turns) = 293mm2.
  • 11 AWG is 2.393mm diameter so its cross-sectional area is 4.498mm2(4.498mm2/turn * 22) turns (total for both windings) = 99mm2.
  • The total of the copper alone is (293 + 99) = 392mm2, not including fill factor.  Using this number with various fill factors we get:
    • Fill factor of 0.3:  392 / 0.3 = 1307mm2150% of the available space - we must do better!
    • Fill factor of 0.4:  392 / 0.4 = 980mm2.  113% of the available space - getting closer.
    • Fill factor of 0.45:  392/0.45 = 871mm2. 100.1% - this is almost exactly how how much room we have.
    • Fill factor of 0.5:  392/0.5 = 784 mm2.  90% - we should be fine if we can do this.
According to this method of calculation we will need to achieve a fill factor of about 0.45 in order to have the turns actually fit. Will the fact that the thin (0.05mm) insulation between layers is thin enough that overlaying windings will take up less "height" if they can fall in the grooves between wires somewhat?  Can this fill factor actually be achieved?

Let's find out.
Figure 2:
The prepared bobbin, at the start of the wind, covered with an initial layer
of polyimide tape.
Click on the image for a larger version.

Winding the transformer:

While it might seem customary to wind the primary first, this is not always the best strategy.  It is often the case that the thinnest wire is wound first as the corners of the bobbin are their sharpest when the diameter is small, making it easy to handle and allowing slightly better packing efficiency and, thus, a better "fill factor."  In this case, because the primary used thinner wire (17 AWG) than the secondary (11 AWG), I wound the primary first.

In preparation for the start of winding I placed a layer of 0.05mm polyimide tape onto the nylon bobbin as a foundation and to give the wire a bit of a surface to "bite" into - and to provide just a little more protection even though it is unlikely that the transformer could ever survive the sorts of conditions that would melt or arc over the bobbin in the first place!

Figure 3:
The three primary voltage taps.
As may be seen, the lower voltage taps (115, 120 volts) consists of a loop
of wire that is brought out of the winding.  The locations of these taps
is staggered somewhat to space apart where they emerge from the side
of the bobbin:  The slight, fractional-turn deviation from the calculated tap
location causes an insignificant voltage change on a winding with this
many turns.  With the taps emerging at a right angle, away from the
"corners" of the bobbin they will add wire height only to the portion
that faces the end bells of the transformers, not on the "sides" between
the winding and the steel laminations which would be on the top
and bottom of this picture.  This method of bring out the taps
also prevents the taps from significantly reducing the number of turns
that will fit on the layer which can keep the number of layers down
to that calculated.
Click on the image for a larger version.
Because 17 AWG wire is actually quite large I "drilled" a hole through the nylon with the conical tip of a hot soldering iron (easier and safer to do than with a drill - particularly when there are already windings present on the bobbin that could be damaged by the bit) and brought the wire straight out the side of the bobbin.  Winding excess length around the screws of the bobbin that were placed there for the purpose of keeping this wire out of the way, I proceeded to place the first layer.

Winding very carefully I laid the turns side-by-side and pushed them closer together to reduce the space after every few turns.  At the end of the first layer I temporarily taped the wire to the side of the bobbin to keep it from unraveling and put an even layer of 0.05mm polyimide tape over the first layer to both insulate and secure the windings before starting the next layer.

Because the first layer was wound very neatly, the second and subsequent layers usually fell into the grooves between the windings of the previous layer with this thin insulating tape which can make it easier to keep these layers nice and neat.  At the ends of the winding there can be a bit of "mechanical confusion" as there is inevitably a sort of "half turn" of spacing between the wire and bobbin that cannot be effectively filled.  As one continues to add layers, this gap on the ends tends to gradually become deeper and care must be taken to make sure that as the wire (inevitably) falls into this gap that it falls atop insulation rather than the previous wire as to minimize the possibility of the wire being chafed and shorted with vibration and thermal cycling.

Figure 4:
A side view of from where the primary taps emerge.  It is important
that the taps be labeled at the time of winding to avoid later
confusion and the possible need to reverse engineer what was done!  Small
pieces of Nomex paper insulation are visible, used to mechanically
separate the overlaying conductors.
Click on the image for a larger version.
At turns 220 and 229 the winding was paused to make the 115 and 120 volt taps.  This was done by making a loop of wire approximately in the middle of the face of the winding, bringing the two wires of the loop together so that they carefully lay side-by-side and bringing it out the side of the bobbin through a hole that was labeled with a permanent marker.  Underneath this loop was placed both some polymide tape and some Nomex (tm) paper insulation to prevent the pressure of the wires of these taps from impinging directly on the insulation of the turns below it and shorting some turns.  At the very end of the winding the tail end of the wire was brought directly out through a labeled hole.

Over the top of the taps was placed an additional layer of polyimide tape and the entire primary was then covered with about 0.5mm of a combination of Nomex paper and polyimide tape to provide a durable insulation between it and the secondary.  Up along the sides of the bobbin a few millimeters of extra insulating tape was added to increase the "creep" distance - an important safety factor when high voltages are concerned.

Once this was done it was time for the secondary windings.  Because 11 AWG is quite large, it takes a bit of brute force to handle.  Using a pair of strong needle-nose a fairly sharp right-angle bend was made in the wire so that it could pass through the slightly oversized hole that I had melted into the side of the bobbin without taking up too much extra space and the winding proceeded with the wire being bent carefully around each corner of the bobbin.

Figure 5:
The completed bobbin, overtaped with taps coming out several sides.
The thin center-tap winding of the outer filament winding (yellow-orange
wire) is easily visible with the purple center tap of the inner filament winding
being seen in the background.  Since the center tap carries only the tubes'
cathode currents, the center taps need only carry a few hundred milliamps
at most.
Click on the image for a larger version.
As it turns out, only about 21 and a fraction turns of the required 22 turns would actually fit across the bobbin so a new layer was started that had just one turn - but this extra turn was lined up with the partial final layer of the primary so its total height was less than it would otherwise have been.  Carefully making a fairly sharp bend in the wire and passing it through a labeled hole in the bobbin, I then located - by counting from each end - the exact location of the 11th turn - the center-tap.  There I carefully scraped the insulation off the top of the wire and, with a very hot soldering iron it was tinned and a short piece of PTFE (Teflon (tm)) covered wire was was attached and brought out through a labeled hole in the side of the bobbin.

While this method of connecting the center-tap is a bit kludgy, the use of magnet wire with a high-temperature polyimide insulation and the underlying polyimide tape between layers minimizes the possibility that the wire itself will be damaged in the process of soldering - and careful visual inspection and tugging on the added tap wire indicated that the connection was quite secure and that the wire itself and insulation in neighboring turns were still intact.  The use of a very hot iron may seem counter-intuitive, but having a lot of heat and thermal mass means that one can thoroughly heat the wire rather quickly to make a proper, alloyed solder connection.  Because the center tap is low-current, needing to carry only a few hundred milliamps of cathode current from the tube, the tap wire is quite small - about 24 AWG.  If I do this technique again I will insert a piece of tape as a "cradle" at the tap point during winding to add extra insulation around the location of the tap and the adjacent turns.

Figure 6:
A side view of the completed bobbin.
Before the transformer's end bells are installed, wires will be attached to
the primary winding's connections and the heavy filament wires will be protected
with an additional layer of insulation where they are brought out.
Click on the image for a larger version.
The first primary completed, it was covered with a layer of 0.05mm polyimide tape, a layer of 0.05mm Nomex paper and another two layers of 0.05mm polyimide tape.

To avoid cluttering the bobbin with too many holes that were too close to each other, the second primary was started nearly 1/4 turn away from the first primary (at nearly the next corner) and since the first had taken a bit more than one layer, I had to "offset" the start of the winding slightly, crossing over the top single-turn top winding of the first primary.  Understandably, this was done with care, bending a slight loop in the wire to go up and over with plenty of insulating tape and a piece of Nomex paper slid underneath to protect the adjacent wires.

The winding proceeded from there, but since it could not start at the end of the bobbin there were now several turns at the end in an "extra" layer that required yet another careful "crossing of the wires" with plenty of insulation.  Upon securing the winding the exact middle of the secondary was located - a task made slightly more difficult by some of the turns being overlaid on a new layer and the center tap was carefully made in the same manner as before.

The winding being done, the second secondary was covered with several layers of polyimide tape and, using a clamp and two pieces of wood, the windings on the two sides of the bobbin that were not facing outwards were squeezed together, slightly reducing the height and increasing the spacing where it passed through the core.

Figure 7:
The "primary side" of the transformer with the solder
joints having been doubly-insulated with heat-
shrinkable tubing.
Click on the image for a larger version.
The results:

As it turned out, the windings - including the unintended partial layer on the secondaries - completely filled up the bobbin, but there was easily a millimeter or two clearance between the windings and the laminations.

In testing the transformer unloaded using a variable transformer I ended up with the following results:
  • 115 volt primary tap at 115.0 volts:  11.49 and 11.48 VAC on windings 1 and 2, respectively
  • 120 volt primary tap at 120.0 volts:  11.49 and 11.49 VAC
  • 125 volt primary tap at 125.0 volts:  11.52 and 11.51 VAC
  • Accuracy of center-tap voltage:  Better than 50 millivolts on each winding.
  • The magnetization current (no load) was approximately 300 mA on each tap at its rated voltage, decreasing somewhat with the higher-voltage taps.  It should be noted that the magnetization current is about 90 degrees out of phase with the reflected load current so it won't count too very much against us when the transformer is actually under load.
Figure 8:
The "secondary" side of the transformer.  The heavy
(11 AWG) wires are first insulated with PTFE
insulation and where they emerge from the metal
bell are covered with colored heat-shrinkable tubing
to identify the windings.
One of the reasons for the primary taps is to allow "fine tuning" of the filament voltage:  Putting taps on the relatively low-current primary is much easier than providing several pairs of equal-spaced taps on the center-tapped, high-current secondary!

As it turns out the actual heater voltage of the tubes that will be used is 10.5 volts, but it is common practice to purposely add a bit of series resistance to reduce the "cold filament" inrush current when the power is first applied - something that will likely involve a drop of a few hundred millivolts through additional resistance:  Anyway, it is much easier to drop a small bit of voltage than add it!

What if we did need more filament voltage than our 11 volt (loaded) target?  The worst-case scenario would be to run the 115 volt tap at 125 volts (yielding 12.5 unloaded volts on the secondaries) which would increase the magnetic flux of the core to an estimated 1.48 Tesla - still within the "safe" range for the M-6 core material!

Lessons learned:

The entire reason for doing this task is to learn something, so here are a few comments:
  • The "tack" method of attaching the center tap wire to the secondaries seems to work OK, but I can see that it was not done very carefully, it could easily go wrong for a number of reasons:
    • Damaging the insulation of adjacent turns and causing immediate or future problems with shorting.  The use of high-temperature polyimide wire allowed this to be done safely, but in the future I would lay the tap point in a "cradle" of polyimide tape to provide additional protection to the adjacent turns.
    • A faulty solder joint due to inadequate breaking of the insulation on the top surface of the wire and/or insufficient heat to make the joint.
    • This method of attaching a comparatively thin conductor is only appropriate where the current through the center tap will be quite small.  In this case, only the cathode current of the tubes - a few hundred milliamps at most - is all that need be conveyed.
  • I did not end up with much additional room on the bobbin when the winding and final insulation layers were completed.  Were I to design and build this transformer again and I were willing to buy whatever sized wire I needed I would probably have used 18 AWG for the primary.
  • 12 AWG would have probably been just fine for the secondaries, particularly with the use of modern, high temperature wire and insulation and the fact that the two secondaries are on the "outside" of the bobbin.  The use of 12 AWG would have also easily allowed each layer of 22 turns to be would with a little bit of room to spare.
  • Had I used 18 and 12 AWG wire for the primary and secondaries, respectively, there would have easily been enough room to add yet another secondary winding such as a 6.3 volt winding for tube filaments or even yet another low-current winding for bias, control logic, or whatever.
Overall, I'm pleased with the results.

Final comments:

The transformer has yet to be encapsulated in insulating varnish and shims have not been inserted between the core laminations and the bobbin, so it hums a lot more now than it will when it is complete.  It will not be until after the initial testing of the (yet to be constructed) amplifier that this will be done as it will still be possible to make slight modifications to the transformer (e.g. change the number of turns, add extra, low-current windings, etc.) in its present state.

In static (no load) testing the transformer was operated with 130 volts applied to the 115 volt tap resulting in an estimated 1.54 Tesla core flux, a 28C (50F) temperature rise was observed.  When 115 volts was applied to the same tap - a situation more representative of core losses (not including the resistive losses in the winding) that might be observed in actual use the temperature rise was just 19C (30F).


How long did it take to wind this thing?  With all of the materials and components lined up it took less than two hours to wind this transformer - being very careful - and about another hour to stack the cores and do initial testing using a variable transformer supply.

The next installment will describe the design and construction of the high voltage plate transformer.

[End]